High Voltage SEPIC Converter

ABSTRACT

A SEPIC converter with over-voltage protection includes a high-side inductor that connects a node V w  to a node V x . The node V x  is connected, in turn to ground by a power MOSFET. The node V x  is also connected to a node V y  by a first capacitor. The node V y  is connected to ground by a low-side inductor. A rectifier diode further connects the node V y  and a node V out  and an output capacitor is connected between the node V out  and ground. A PWM control circuit is connected to drive the power MOSFET. An over-voltage protection MOSFET connects an input supply to the PWM control circuit and the node V w . A comparator monitors the voltage of the input supply. If that voltage exceeds a predetermined value Vref the comparator output causes the over-voltage protection MOSFET to disconnect the node V w  and the PWM control circuit from the input supply.

This application is a continuation of application Ser. No. 11/933,402, filed Oct. 31, 2007, now U.S. Pat. No. 8,350,546, issued Jan. 8, 2013, which is incorporated herein by reference in its entirety.

BACKGROUND OF THE INVENTION

Most DC to DC switching voltage regulators such as the Buck converter and boost converter are capable of only regulating a voltage above or below a given input but not capable of both step up and step down regulation. A SEPIC (single ended primary inductor converter) is a DC-DC converter which allows the output voltage to be greater than, less than, or equal to the input voltage. The output voltage of the SEPIC is controlled by the duty cycle of the control transistor. The largest advantage of a SEPIC over the buck-boost converter is a non-inverted output (positive voltage). SEPICs are useful in applications where the battery voltage can be above and below the regulator output voltage. For example, a single Lithium ion battery typically has an output voltage ranging from 4.2 Volts to 3 Volts. If the accompanying device requires 3.3 Volts, then the SEPIC would be effective since the battery voltage can be both above and below the regulator output voltage. Other advantages of SEPICs are input/output isolation and true shutdown mode: when the switch is turned off output drops to 0 V.

As shown in FIG. 1 a prior art SEPIC converter 1 comprises a PWM control circuit 2, N-channel power MOSFET 3 with intrinsic drain-to-source diode 4, high-side inductor 5 capacitor 6, low-side inductor 7, rectifier diode 8 and output capacitor 9 powering load 10. Operation comprises repeatedly magnetizing inductor 5 whenever MOSFET 3 is in its ON and conducting state and transfer ring energy to output capacitor 9 and load 10 in alternating phases.

During operation, the node voltage V_(x) peaks at a voltage (V1N+VouT). The BVoss breakdown of MOSFET 3 and diode 4 must exceed this peak voltage with some reserve.

The converter as shown cannot survive an aver-voltage condition because no means exists to stop the switching operation of MOSFET 3. Instead of limiting the maximum input voltage, converter I continues to operate at any input voltage until the drain voltage on MOSFET 3 exceeds safe limits and damages the device. In addition to this inability to survive high input voltages, PWM controller 2 contains low-voltage control circuitry which cannot operate when powered directly from a high voltage input.

The circuit as shown also suffers from the lack of a true load disconnect. Current sensing is also problematic since there is no convenient means to detect the input current in inductor 5.

What is needed is a SEPIC converter offering high-voltage operation up to some safe level below the rating of the power MOSFET, a means to inhibit switching operation under excessive input voltage conditions, the ability to disconnect the load from the input, and a means to detect the input current either to implement current mode control, to prevent over-current conditions, or ideally both.

SUMMARY OF THE INVENTION

The present invention provides a family of SEPIC converters that overcome the disadvantages of the prior art. A basic building block of this family is the generic SEPIC converter shown in FIG. 1. This converter includes a high-side inductor that connects a node V_(w) to a node V_(x). The node V_(x) is connected, in turn to ground by a power MOSFET. The node V_(x) is also connected to a node V_(v) by a first capacitor. The node V_(v) is connected to ground by a low-side inductor, A rectifier diode further connects the node V_(v) and a node V_(out) and an output capacitor is connected between the node V_(out) and ground.

A PWM control circuit is connected to drive the power MOSFET. The PWM control circuit turns the power MOSFET ON an OFF in a repeating pulse-width-modulation pattern. The duty cycle of the power MOSFET is varied in proportion to the voltage at the node V_(out) to maintain the output of the SEPIC converter within regulation. Whenever the power MOSFET is ON, current from the input supply magnetizes the high-side inductor. When the power MOSFET subsequently turns OFF, the energy stored in the magnetic field of the inductor is transferred to the output capacitor. A load connected over the output capacitor is powered in this fashion.

To this basic SEPIC topology just described, a first embodiment of the present invention adds circuitry for over-voltage protection. Specifically, this embodiment uses over-voltage protection MOSFET to connect the input supply to the PWM control circuit and the node V_(w). The over-voltage protection MOSFET is driven by the output of a comparator. The comparator is connected to monitor the difference between a predetermined reference voltage v_(ref) and the voltage of the input supply. If the voltage of the input supply exceeds the predetermined value v_(ref) the comparator output causes the over-voltage protection MOSFET to disconnect the node V_(w) and the PWM control circuit from the input supply, In this way, a SEPIC converter is provided that can survive input voltages that would otherwise damage the power MOSFET or PWM control circuit.

A second embodiment of the present invention provides a high-voltage SEPIC converter. For this embodiment (once again, based on the topology described above) a linear regulator (typically, an LDO) is used to supply current from the input supply to the PWM control circuit. In this way, the voltage applied to the PWM control circuit never exceeds the regulated output of the linear regulator and the SEPIC converter can survive input voltages that would otherwise damage the PWM control circuit.

A third embodiment of the present invention provides a high-voltage SEPIC with over-voltage protection. This embodiment includes all of the elements of the high-voltage SEPIC just described. To add over-voltage protection, an AND gate is added to drive the power MOSFET. The inputs to the AND gate are the output of the PWM control circuit and the output of a comparator. The comparator is connected to monitor the difference between a predetermined reference voltage v_(ref) and the voltage of the input supply. If the voltage of the input supply exceeds the predetermined value v_(ref) the comparator output causes the AND gate to disable the output of the PWM control circuit. In turn, this causes the power MOSFET to turn OFF.

By shutting down the power MOSFET under over-voltage conditions, the SEPIC converter ensures that the drain of the power MOSFET never exceeds the voltage of the input supply. In this way, the power MOSFET is protected in over-voltage conditions. At the same time, the PWM circuit protected by the LDO as described for the previous embodiment.

A fourth embodiment of the present invention provides a high-voltage SEPIC with over-voltage protection and load disconnect. This embodiment includes all of the elements of the high-voltage SEPIC with over-voltage protection just described. To add load disconnect, current sensing and load disconnect circuitry is added to monitor the current flowing from the input supply to the high-side inductor. The current sensing and load disconnect circuitry includes a P-channel MOSFET connected between the input supply and the high-side inductor. During normal operation, this P-channel device is biased to allow current to pass to the high-side inductor.

The current through the P-channel MOSFET is monitored using a current mirroring technique. This produces an over-current signal that is connected as an input to the AND gate (in this implementation the AND gate has three inputs) if current flowing through the P-channel MOSFET exceeds a predetermined value, the over-current signal causes the AND gate to disable the output of the PWM control circuit. In turn, this causes the power MOSFET to turn OFF and protects the SEPIC converter from over-current damage. The use of a P-channel MOSFET to monitor the current to the high-side inductor also means that, by biasing that MOSFET to be OFF, the SEPIC converter may be disconnected from the input supply.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic of a conventional SEPIC converter (Prior Art).

FIG. 2 is a schematic of a SEPIC converter with series over-voltage protection as provided by an embodiment of the present invention.

FIG. 3 is a schematic of a high voltage SEPIC without over-voltage protection as provided by an embodiment of the present invention.

FIG. 4 is a schematic of a high voltage SEPIC with over-voltage protection as provided by an embodiment of the present invention.

FIG. 5 is a schematic of a high voltage SEPIC with over-voltage protection, current sensing, over-current protection and true load disconnect as provided by an embodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Over-Voltage-Protected SEPIC

One means to extend the voltage range of the SEPIC converter is to utilize an over-voltage protection, i.e. OVP, circuit that disconnects the converter from the input in the event the input voltage exceeds a pre-specified value. In converter 20 of FIG. 2, SEPIC converter 21 is protected by P-channel MOSFET 31 controller by comparator 33 which compares the input voltage V1N to a reference voltage Vref. Reference voltage 34 may be implemented using a bandgap reference, Zener diode, a series of forward biased diodes or any other well known voltage reference technique, or a scaled multiple of said voltage. P-channel MOSFET 31 includes reverse-biased intrinsic P-N diode 32 with its cathode tied to V1N and its anode connected to the input to converter 21.

SEPIC converter 21 comprises a PWM control circuit 22, N-channel power MOSFET 23 with intrinsic drain-to-source diode 24, high-side inductor 25, capacitor 26, low-side inductor 27, rectifier diode 28 and output capacitor 29 powering load 30. Operation comprises repeatedly magnetizing inductor 25 whenever MOSFET 23 is in its ON and conducting state and transferring energy to output capacitor 29 and load 30 in alternating phases.

Whenever V1N is below Vref the gate of P-channel 31 is pulled down by comparator 33 and P-channel 31 is turned on. The maximum gate to source voltage VcsP cannot exceed the maximum gate rating of the P-channel 31, i.e. VcsP<I V1N−VcpI. Accordingly, the Vee input of SEPIC converter 21 is connected to VIN and the converter is operating. Whenever V1N is above Vref an over-voltage condition has occurred and the input Vee of SEPIC converter 21 is disconnected from V1N and allowed to float or alternatively is grounded. The Vref voltage determines the maximum value of Vee powering SEPIC converter 21 and PWM controller 22. PWM controller must therefore utilize devices capable of operating at the maximum allowed We voltage, i.e. Vref. During operation, the node voltage V_(x) peaks at a voltage (V1N+VouT)<(Vref+VouT). The BVoss breakdown of MOSFET 23 and diode 24 lust exceed this peak voltage with some guardband.

As such OVP protection MOSFET 31 protects SEPIC converter 21 but must be rated for the maximum V1N input voltage. The devices used in PWM control circuit 22 must support the same voltage rating. An even higher voltage is imposed on node V_(x) and across diode 24 of N-channel MOSFET 23. Converter 20 therefore requires two high voltage MOSFETs, N-channel 23 and P-channel 31 with respective on-resistances RosN and RosP, to implement an over-voltage protected SEPIC converter. The current capability of the converter is adversely affected by the higher on-resistance of such high voltage devices, i.e. Rtotal=(RosP+RosN). While the approach of converter 20 may be used for any input voltage, practically these considerations limit the input voltage to the 12V to 18V range, especially in implementing PWM controller 22.

Another benefit of OVP protected SEPIC converter 20 is its ability to implement the load-disconnect function, simply by turning OFF P-channel MOSFET 31 by biasing its gate to its source potential, Le, where VcP=V1N.

High-Voltage SEPIC

Another method to extend the voltage range of the SEPIC converter is to utilize a high voltage low-drop-out linear regulator, or LDO, to protect the control circuitry from high voltages up to a pre-specified value. In SEPIC converter 50 of FIG. 3 linear regulator 60 limits the maximum voltage imposed on PWM controller 51 to some predefined maximum voltage Vee, typically 3V or 5V, so that the devices utilized within PWM circuit 51 may comprise only low-voltage devices.

SEPIC converter 50 includes a PWM control circuit 51, N-channel power MOSFET 52 with intrinsic drain-to-source diode 53, high-side inductor 54, capacitor 55, low-side inductor 56, rectifier diode 57 and output capacitor 58 powering load 59. Operation comprises repeatedly magnetizing inductor 54 whenever MOSFET 52 is in its ON and conducting state and transferring energy to output capacitor 58 and load 59 in alternating phases.

As illustrated linear regulator 60 is preferably a low-drop-out type, e.g. with a series voltage drop under 200 mV, to extend the operating voltage range of converter 50 to lower input voltage V1N. The design of low drop-out linear regulators is well known to one skilled in the art of power electronics. Input and output capacitors 61 and 62 act as filter capacitors and prevent LDO 60 from oscillating. The benefit of the smaller sized devices is the silicon die area may be reduced compared to the area occupied by high voltage PWM circuit 22 of aforementioned SEPIC converter 20.

While LDO 60 protects PWM controller 51 from high input voltages it does not limit the voltage on the remainder of the converter circuit or on MOSFET 52. Lacking any over-voltage protection circuitry and series disconnect switch, the N-channel MOSFET 52 and diode 53 must be rated to operate up to the maximum input voltage with adequate guard-banding to avoid accidental or momentary avalanche breakdown. During operation the peak V_(x) voltage is typically (V1N+VouT).

Since only one high voltage MOSFET is present in converter 50, the current capability of the converter is improved in comparison to converter 20, since the total MOSFET resistance is only that of MOSFET 52, i.e. where Rtotal=(RosN). Without over-voltage protection however, the breakdown voltage of MOSFET 52 must be higher than (V1N+VouT). In this approach, the breakdown voltage guard-band of MOSFET 52 increases with increasing input voltage. While the approach of converter 50 may be used for any input voltage, practically these considerations preferably limit the input voltage to the 18V to 30V range, beyond which the need for excessive voltage guard-banding makes MOSFET 52 unnecessarily large to compensate for its higher resistance.

Without OVP protection, one disadvantage of high-voltage SEPIC converter 50 is its inability to offer the load-disconnect function. As a result the circuit provides no means to disconnect load 59 from V1N.

High-Voltage SEPIC With Over-Voltage Protection

An improved SEPIC converter combines the over-voltage protection features of converter 20 with the high voltage capability of converter 50. The resulting OVP protected high-voltage SEPIC converter is illustrated in converter 70 of FIG. 4. As such, a comparator 83 in conjunction with a high voltage MOSFET 72 protects the power circuitry while a linear regulator protects the PWM control circuit 69 from high voltages. Unlike in converter 20, OVP protection is achieved without inserting a second high voltage device in the high current path, but instead is achieved by changing the control of the high-voltage rated low-side N-channel MOSFET 72.

Specifically, linear regulator 80 limits the maximum voltage imposed on PWM controller 69 to some predefined maximum voltage Vee, typically 3V or 5V, so that the devices utilized within PWM controller 69 may comprise only low-voltage devices. Linear regulator 80 is preferably a low-drop-out type, e.g. with a series voltage drop under 200 mV, to extend the operating voltage range of converter 70 to lower input voltages V1N. The design of low drop-out linear regulators is well known to one skilled in the art of power electronics. Input and output capacitors 81 and 82 act as filter capacitors and prevent LDO 80 from oscillating. The benefit of the smaller sized devices is the silicon die area may be reduced compared to the area occupied by high voltage PWM circuit 22 of aforementioned SEPIC converter circuit 20.

For over-voltage protection SEPIC converter 70 achieves OVP capability by shutting OFF high-voltage MOSFET 72 whenever an over-voltage condition is detected. The maximum operating voltage of high voltage N-channel MOSFET 72 is set by OVP reference voltage 84. Comparing the input voltage V1N to the reference voltage Vref, at the point of over-voltage shutdown when V1N>Vref, comparator 83 inhibits PWM control of MOSFET 72 with logic AND gate 71. in such an event PWM controller 69 no longer determines the turn ON and OFF of MOSFET 72. The maximum V_(x) voltage at this moment is (Vref+VouT) plus some guard banding. Above this voltage, converter 70 no longer functions and the maximum voltage imposed on the drain of OFF state MOSFET 72 is simply VIN.

So in improved SEPIC converter 70, the voltage capability of MOSFET 72 and diode 73 needed for operation is used to achieve the OVP function without adding extra series resistance to the high-current power path. Comparator 83 is used to monitor the input voltage V1N and compare it to an over-voltage reference set to a voltage Vref. Reference voltage 84 may be implemented using a bandgap reference, Zener diode, a series of forward biased diodes or any other well known voltage reference technique, or a scaled multiple of said voltage.

The remaining elements of SEPIC converter 70 comprises a PWM control circuit 69, N-channel power MOSFET 72 with intrinsic drain-to-source diode 73, high-side inductor 74, capacitor 75, low-side inductor 76, rectifier diode 77 and output capacitor 78 powering load 79. Normal operation comprises repeatedly magnetizing inductor 74 whenever MOSFET 72 is in its ON and conducting state and transferring energy to output capacitor 78 and load 79 in alternating phases.

Since only one high voltage MOSFET is present in converter 70, the current capability of the converter is improved in comparison to converter 20, since the total MOSFET resistance is only that of MOSFET 72, i.e. where Rtotal=(RosN). With over-voltage protection, the breakdown voltage of MOSFET 72 must be only slightly higher than (Vref+VouT) offering the need for less voltage guard banding and on-resistance penalty. Therefore, the approach of converter 70 may be used for any input voltage with minimal impact on conversion efficiency.

Like converter 50, without a series P-channel MOSFET, OVP protected high-voltage SEPIC converter 70 is unable to offer the load-disconnect function. As a result the circuit provides no means to disconnect load 79 from V1N.

High-Voltage SEPIC With Over-Voltage Protection And Load Disconnect

In another embodiment of this invention, an improved SEPIC converter combines the over-voltage protection features and load disconnect capability of converter 20 with the high voltage capability of converter 50. The resulting OVP protected high-voltage SEPIC converter 90 is illustrated in FIG. 5 including current sensing and load disconnect circuitry 106.

As such, an over-voltage protection comparator 104 in conjunction with a high voltage MOSFET 93 protects the power circuitry while a linear regulator 102 protects the PWM control circuit 91 from high voltages. Similar to converter 70 and unlike in converter 20, OVP protection is achieved without inserting a second high voltage device in the high current path, but instead is achieved by changing the control of the high-voltage rated low-side N-channel MOSFET 93.

To avoid the need for substantial high voltage circuitry, linear regulator 102 limits the maximum voltage imposed on PWM controller 91 to some predefined maximum voltage Vee, typically 3V or SV, so that the devices utilized within PWM circuit 91 may comprise only lover-voltage devices. Linear regulator 102 is preferably a low-drop-out type, e.g. with a series voltage drop under 200 mV, to extend the operating voltage range of converter 90 to lower input voltages V1N. The design of low drop-out linear regulators is well known to one skilled in the art of power electronics. Input and output capacitors 101 and 103 act as filter capacitors and prevent LDO 102 from oscillating. The benefit of the smaller sized devices is the silicon die area may be reduced compared to the area occupied by high voltage PWM circuit 22 of aforementioned SEPIC converter circuit 20.

For over-voltage protection SEPIC converter 90 achieves OVP capability by shutting OFF high-voltage MOSFET 93 whenever an over-voltage condition is detected. The maximum operating voltage of high voltage N-channel MOSFET 93 is set by OVP reference voltage 105. Comparing the input voltage V1N to the reference voltage Vref, at the point of over-voltage shutdown when V1N>Vref, comparator 104 inhibits PWM control of MOSFET 93 with triple-input logic AND gate 92. In such an event PWM controller 91 no longer determines the turn ON and OFF of MOSFET 93. The maximum V_(x) voltage at this moment is (Vref+VouT) plus some guard banding. Above this voltage, converter 90 no longer functions and the maximum voltage imposed on the drain of OFF state MOSFET 93 is simply V1N.

In improved SEPIC converter 90 the voltage capability of MOSFET 93 and diode 94 needed for operation is used to achieve the OVP function without adding extra series resistance to the high-current power path. Comparator 104 is used to monitor the input voltage V1N and compare it to an over-voltage reference set to a voltage Vref. Reference voltage 105 may be implemented using a bandgap reference, Zener diode, a series of forward biased diodes or any other well known voltage reference technique, or a scaled multiple of said voltage.

Current sensing is achieved in improved SEPIC converter 90 using current sensing and load disconnect circuitry 106, utilizing a low-loss current sensing technique described in a pending U.S. patent application entitled “Cascade Current Sensor for Discrete Power Semiconductor Devices” by R. K. Williams. That disclosure is incorporated in this document by reference. Rather than using a resistor as a current sense element, low-voltage low-resistance P-channel MOSFET 107 with intrinsic reverse biased P-N diode 108 is inserted in the path of the input current flowing in inductor 95. Under normal operation the gate voltage Vc_(p) of P-channel MOSFET 107 is pulled down by gate buffer 109 to fully enhance the MOSFET 107 into a low-resistance state with a resistance RosP for a given area substantially lower than that of high-voltage P-channel MOSFET 31 described previously in FIG. 2.

The maximum gate to source voltage Vcs_(p) of P-channel MOSFET 107 in its ON condition cannot exceed the maximum gate rating of the P-channel MOSFET 107, i.e. Vcs_(p)<IV_(IN)−Vc_(p)I as determined by the output of gate buffer 109. When MOSFET 107 is ON and conducting, amplifier or comparator 110 is used to determine the input current flowing into inductor 95. By using a mirror technique the current in MOSFET 107 can be accurately determined.

During normal operation, the gate buffer 109 biases MOSFET 107 into a low-resistance conducting state. Amplifier or comparator 110 accurately detects the current flowing in conducting MOSFET 107 and outputs a signal. If this signal is analog, representing a measurement of the current in inductor 95, the information may be used to implement current mode control of PWM controller 91.

In another implementation of the embodiment shown in FIG. 5, comparator 110 has a digital output representing over-current protecting shutdown or OCS, and used as one input to triple NAND gate 110. Only when converter 90 has an input voltage VIN below a specified preset level and the measured current in MOSFET 107 does not cause comparator 110 to flip states as an over-current condition, then the output of PWM controller 91 controls the turning ON and OFF of N-channel MOSFET 93. Accordingly, the V_(w) input of SEPIC converter 90 is connected to V1N and the converter is operating.

The remaining elements of SEPIC converter 90 comprise N-channel power MOSFET 93 with intrinsic drain-to-source diode 94, high-side inductor 95, capacitor 96, low-side inductor 97, rectifier diode 98 and output capacitor 99 powering load 100. Normal operation comprises repeatedly magnetizing inductor 95 whenever MOSFET 93 is in its ON and conducting state and transferring energy to output capacitor 99 and load 100 in alternating phases.

Since only one high voltage MOSFET 93 plus one low-voltage MOSFET 107 is present in converter 90, the current capability of the converter is improved in comparison to converter 20, since the total MOSFET resistance is that of high-voltage MOSFET 93 plus the resistance of low-voltage MOSFET 107, i.e. where R_(total)=(RosP+RosN). With over voltage protection, the breakdown voltage of MOSFET 93 must be only slightly higher than (Vref+VouT) offering the need for less voltage guard banding and on-resistance penalty. Therefore, the approach of converter 90 may be used for any input voltage with minimal impact on conversion efficiency.

By including series low voltage P-channel MOSFET 107, the disclosed OVP protected high-voltage SEPIC converter 90 is able to offer the load-disconnect function whereby the circuit provides a means to disconnect load 100 from V1N Load disconnect is controlled by P-type current sense PCS signal, the input to gate buffer 109. 

1. A single ended primary inductor converter with over-voltage protection comprising: an over-voltage protection MOSFET connected to an input terminal of the converter; and a comparator connected to drive a gate of the over-voltage protection MOSFET so as to turn off the over-voltage protection MOSFET and thereby disconnect the input terminal when a voltage at the input terminal exceeds a predetermined value.
 2. A single ended primary inductor converter with over-voltage protection comprising: a high-side inductor connected in series with a power MOSFET; a PWM control circuit connected to drive a gate of the power MOSFET; and a linear regulator connected to control a flow of current from an input terminal of the converter to the PWM control circuit.
 3. The single ended primary inductor converter of claim 2 further comprising a logic circuit connected between the PWM control circuit and the gate of the power MOSFET for preventing the PWM control circuit from driving the power MOSFET when a voltage at the input terminal exceeds a predetermined value.
 4. The single ended primary inductor converter of claim 3 further comprising a current sensing circuit configured to generate an over-current signal as a function of the magnitude of a current in the high-side inductor, the current sensing circuit being connected to the logic circuit.
 5. The single ended primary inductor converter of claim 4 wherein the current sensing circuit comprises: a MOSFET connected between the input terminal and the high-side inductor; and a comparator having input terminals connected to source and drain terminals, respectively, of the MOSFET and an output terminal connected to the logic circuit.
 6. A method of controlling a single ended primary inductor converter, comprising disconnecting an input terminal of the converter from a high-side inductor when the voltage at the input terminal exceeds a predetermined value.
 7. A method of controlling a single ended primary inductor converter, the converter comprising a high-side inductor connected in series with a power MOSFET, and a PWM control circuit connected to drive a gate of the power MOSFET, the method comprising controlling a flow of current from an input terminal of the converter to the PWM control circuit.
 8. The method of claim 7 further comprising: generating an over-voltage signal as a function of a voltage at the input terminal; and preventing the PWM control circuit from driving the power MOSFET when the over-voltage signal indicates that the voltage at the input terminal has exceeded a predetermined value.
 9. The method of claim 8 further comprising: generating an over-current signal as a function of the magnitude of a current in the high-side inductor; and preventing the PWM control circuit from driving the power MOSFET when the over-current signal indicates that the magnitude of the current in the high-side inductor has exceeded a predetermined value.
 10. The method of claim 8 wherein generating the over-current signal comprises: connecting a MOSFET between the input terminal and the high-side inductor; biasing the MOSFET so that a current flows through the MOSFET from the input terminal to the high-side inductor; and generating the over-current signal as a function of a voltage drop across the MOSFET.
 11. The single ended primary inductor converter of claim 1 further comprising: a high-side inductor connected between a node V_(w) and a node V_(x); a power MOSFET connected in series between the node V_(x) and ground; a first capacitor connected between the node V_(x) and a node V_(y); a low-side inductor connected between the node V_(y) and ground; a rectifier diode connected between the node V_(v) and a node V_(out); and a PWM control circuit connected to drive the gate of the power MOSFET.
 12. The single ended primary inductor converter of claim 2 wherein the high-side inductor and the power MOSFET are connected together at a node V_(x) and the power MOSFET is connected between the node V_(x) and ground, the converter further comprising: a first capacitor connected between the node V_(x) and a node V_(v); a low-side inductor connected between the node V_(y) and ground; and a rectifier diode connected between the node V_(v) and a node V_(out).
 13. The single ended primary inductor converter of claim 3 wherein the logic circuit comprises: an AND gate connected between the PWM control circuit and the gate of the power MOSFET; and a comparator for generating an over-voltage signal as a function of the voltage of the input terminal, the over-voltage signal being delivered to an input of the AND gate so that the AND gate prevents the PWM control circuit from driving the power MOSFET when the voltage at the input terminal exceeds the predetermined value.
 14. The single ended primary inductor converter of claim 5 wherein the logic circuit comprises an AND gate connected between the PWM control circuit and the gate of the power MOSFET, an output terminal of the comparator being connected to an input term in of the AND gate.
 15. The method of claim 6 wherein the high-side inductor is connected between a node V_(w) and a node V_(x).
 16. The method of claim 15 wherein the converter further comprises a power MOSFET connected in series between the node V_(x) and ground, a first capacitor connected between the node V_(x) and a node V_(y), a low-side inductor connected between the node V_(y) and ground, a rectifier diode connected between the node V_(y) and a node V_(out), and a PWM control circuit connected to drive the gate of the power MOSFET.
 17. The method of claim 7 wherein controlling the flow of current from the input terminal of the convener to the PWM control circuit comprising using a linear regulator.
 18. The method of claim 7 wherein the high-side inductor is connected between a node V_(w) and a node V_(x) and the power MOSFET is connected between the node V_(x) and ground.
 19. The method of claim 18 wherein the converter further comprises a first capacitor connected between the node V_(x) and a node V_(y); a low-side inductor connected between the node V_(y) and ground; and a rectifier diode connected between the node V_(y) and a node V_(out).
 20. A single ended primary inductor converter comprising: high-side inductor connected to an input terminal and in series with a power MOSFET; a capacitor and a rectifier diode connected in series to a common node between the high-side inductor and the power MOSFET; a low-side inductor connected to a common node between the capacitor and the rectifier; means for detecting a difference between a voltage at the input terminal and a reference voltage; and means for preventing a current from flowing in the high-side inductor responsive to an output signal from the means for detecting.
 21. The single ended primary inductor converter of claim 20 further comprising a PWM control circuit for driving a gate of the power MOSFET and means for preventing a current from flowing to the PWM control circuit responsive to the output signal from the means for detecting.
 22. A single ended primary inductor converter comprising: a high-side inductor connected to an input terminal and in series with a power MOSFET; a PWM control circuit for driving a gate of the power MOSFET; a capacitor and a rectifier diode connected in series from common node between the high-side inductor and the power MOSFET; a low-side inductor connected to a common node between the capacitor and the rectifier; and means for preventing a current from flowing to the PWM circuit responsive to the magnitude of a voltage at the input terminal.
 23. A single ended primary inductor converter comprising: a high-side inductor connected to an input terminal and in series with power MOSFET; a PWM control circuit for driving a gate of the power MOSFET; a capacitor and a rectifier diode connected in series from common node between the high-side inductor and the power MOSFET; a low-side inductor connected to a common node between the capacitor and the rectifier; and means for preventing the PWM control circuit from driving the gate of the power MOSFET responsive to a predetermined condition.
 24. The single ended primary inductor converter of claim 23 wherein the means for preventing the PWM control circuit from driving the gate of the power MOSFET is responsive to the magnitude of a voltage at the input terminal.
 25. The single ended primary inductor converter of claim 23 wherein the means for preventing the PWM control circuit from driving the gate of the power MOSFET is responsive to the magnitude of a current in the high-side inductor.
 26. The single ended primary inductor converter of claim 25 wherein the means for preventing the PWM control circuit from driving the gate of the power MOSFET is also responsive to the magnitude of a voltage at the input terminal. 